1 Recommendation ITU-R M.1177-4 (04/2011) Techniques for measurement of unwanted emissions of radar systems M Series Mobile, radiodetermination, amateur and related satellite services
2 ii Rec. ITU-R M.1177-4 Foreword The role of the Radiocommunication Sector is to ensure th e rational, equitable, efficien t and economical use of the including satellite services, and carry out studies without radio-frequency spectrum by all radiocommunication services, limit of frequency range on the basis of which Recommendations are adopted. The regulatory and policy functions of the Radiocommunication Sector are performed by World and Regional Radiocommunication Conferences and Radiocommunication Assemblies supported by Study Groups. Policy on Intellectual Property Right (IPR) ITU-R policy on IPR is described in the Common Patent Po licy for ITU-T/ITU-R/ISO/IEC referenced in Annex 1 of Resolution ITU-R 1. Forms to be used for the submission of patent statements and licensing declarations by patent http://www.itu.int/ITU-R/go/patents/en where the Guidelines for Implementation of the holders are available from Common Patent Policy for ITU-T/ITU-R/ISO/IEC and the ITU-R patent information database can also be found. Series of ITU-R Recommendations http://www.itu.int/publ/R-REC/en (Also available online at ) Title Series Satellite delivery BO Recording for production, archival and play-out; film for television BR Broadcasting service (sound) BS Broadcasting service (television) BT Fixed service F M Mobile, radiodetermination, amateu r and related satellite services Radiowave propagation P Radio astronomy RA Remote sensing systems RS Fixed-satellite service S Space applications and meteorology SA Frequency sharing and coordination between fixed-satellite and fixed service systems SF Spectrum management SM Satellite news gathering SNG Time signals and frequency standards emissions TF Vocabulary and related subjects V Note This ITU-R Recommendation was approved in English un der the procedure detailed in Resolution ITU-R 1. : Electronic Publication Geneva, 2011 ITU 2011 All rights reserved. No part of this publ ication may be reprodu ced, by any means whatsoever, wi thout written permission of ITU.
3 Rec. ITU-R M.1177-4 1 * RECOMMENDATION ITU-R M.1177-4 Techniques for measurement of unwa nted emissions of radar systems (Question ITU-R 202/5) (1995-1997-2000-2003-2011) Scope t of radiated radar unwanted This Recommendation provides two techniques for the measuremen ious domain emissions and to check emission emissions. It should be used to measure the spur 3 (Section II) of the Radio Regulations (RR), or to power against limits specified in Appendix falling within the out-of-band domain. measure the level of unwanted emissions The ITU Radiocommunication Assembly, considering that both fixed and mobile radar stations in the radiod etermination service are widely a) implemented in bands adjacent to and in harmonic relationship with other services; b) ble to interference from radar stations with that stations in other services are vulnera unwanted emissions with high peak power levels; c) that many services have adopted or are pl anning to adopt digital modulation systems which are more susceptible to interference from radar unwanted emissions; d) that under the conditions stated in considering a) through c), interferen ce to stations in other services may be caused by a radar station with un wanted emissions with high peak power levels; that Recommendation ITU-R SM.329 specifies the maximum values of unwanted e) emissions in the spurious emission domain from radio transmitters; f) that Recommendation ITU-R SM.1541 specifies the generic limits for unwanted emissions in the out-of-band domain, recommends that measurement techniques as described in 1 Annex 1 should be used to provide guidance in quantifying radiated unwanted emission levels from radar stations op erating above 400 MHz; 2 that measurement techniques as described in either Annex 1 or Anne x 2 should be used, as appropriate based upon radar design , to provide guidance in measur ing radiated unwanted emission levels for radar stations opera ting between 50 MHz and 400 MHz; 3 that measurement techniques described in Anne x 2 should be used to provide guidance in quantifying radiated unwanted emission levels from radar stations operating below 50 MHz. * This Recommendation should be brought to the atte ntion of the International Maritime Organization (IMO), the International Civil Aviation Organiza tion (ICAO), the International Maritime Radio Association (CIRM), the World Meteorological Or ganization (WMO) and Ra diocommunication Study Groups 1 and 4.
4 2 Rec. ITU-R M.1177-4 Annex 1 Measurement of unwanted emi ssions of radar systems 1 and 2 recommends as detailed in 1 Introduction Two measurement techniques known as the dire ct and indirect met hods are described. The direct measurement method is recommended a nd measures unwanted emissions from all radars including those that preclude measurements at intermediate points within the radar transmitters. Examples include those which use distributed-transmitter arrays built into (or comprising) the antenna structure. nts of the radar and then combines the results. The indirect method separately measures the compone The recommended split of the radar is to separate the system after the Rotating Joint (Ro-Jo) and thus to measure the transmitter output spectrum at the output port of the Ro-Jo and to combine it with the measured antenna gain characteristics. 2 Reference bandwidth B For radar systems, the reference bandwidth, , used to define unwanted emission limits ref and RR Appendix 3) s (Recommendations ITU-R SM.329 and ITU-R SM.1541, hould be calculated for each particular radar system. For the four ge neral types of radar pulse modulation utilized for radionavigation, radiolocation, ac quisition, tracking and other radiodetermination functions, the reference bandwidth values are determined using the following formulas: for fixed-frequency, non-pulse-coded radar, one divided by the radar pulse length (e.g. if – 1 MHz); μ μ s = s, then the reference bandwidth is 1/1 the radar pulse length is 1 – for fixed-frequency, phase-coded pulsed ra dar, one divided by th e phase chip length (e.g. if the phase coded chip is 2 μ s long, then the reference bandwidth is 1/2 μ s 500 kHz); = for FM or chirped radar, the square root by dividing the chirp – of the quantity obtained μ s) (e.g. if the FM is from 1 250 MHz to 1 280 bandwidth (MHz) by the pulse length ( MHz or 30 MHz during the pulse of 10 s, then the reference bandwidth is μ 1/2 = 1.73 MHz); μ (30 MHz/10 s) – for radars operating with multiple waveforms the reference bandwidth is determined empirically from observations of the ra dar emission. The empirical observation is receiver is tuned to one of the fundamental performed as follows: the measurement system frequencies of the radar, or is tuned to the centre frequency within the chirp range of the radar. The measurement system bandwidth is lable value, and the set to the widest avai received power level from the radar in this bandwidth is recorded. The measurement bandwidth is then progressively narrowed, and the received power level is recorded as a function of the bandwidth. The e nd result is a graph or table showing measured power as a function of measurement system bandwidth. The required bandwidth is the smallest bandwidth in which the full power level is st ill observed and the reference bandwidth can be calculated from a knowledge of the impulse response of the measurement receiver using the factor, measurement bandwidth ratio (MBR ), as described below. If a reduction in power level is observed immediately, then the widest available bandwidth should be used.
5 Rec. ITU-R M.1177-4 3 ter than 1 MHz, then a reference bandwidth, In all cases, where the bandwidths above are grea B , ref of 1 MHz should be used. Measurement bandwidth and detector parameters 3 The measurement bandwidth, B , is defined as the impulse bandwidth of the receiver and is greater m B than the IF bandwidth, , (sometimes referred to as resolution bandwidth for spectrum analyzers). if B , may be derived from the following equation: The measurement bandwidth, m B B × = MBR m if The MBR needs to be determined for the measurement receiver being used. MBR is approximately 3/2 for a –3 dB IF bandwidth Gaussian filter as typically used in many commercial spectrum analyzer receivers (in some instruments the IF bandwidth is defined at the –6 dB point). An appropriate receiver IF bandwid th should be selected to give one of the following recommended measurement bandwidths. (1/ T Measurement T ) for fixed-frequency, non- pulse-coded radars, where is the pulse ≤ 1 B bandwidth length (e.g. if the radar pulse length is 1 μ s, then the measurement m ≤ 1/(1 bandwidth should be = s) = 1 MHz). μ is the phase- (1/ t ) for fixed-frequency, phase-c oded pulsed radars, where t ≤ μ s pulses, each pulse consisting o chip length (e.g. if the radar transmits 26 f 13 phase coded chips that are 2 μ s in length, then the measuremen t bandwidth should be ≤ 1/(2 μ s) = 500 kHz). 1 / 2 / T ) B ( for swept-frequency (FM, or chirp) radars, where B f is the range o ≤ c c T is the pulse le ngth (e.g. if rada r frequency sweep during each pulse and sweeps (chirps) across the frequency range of 1 280 MHz ( = 30 MHz 250-1 and if the pulse length is 10 μ s, then the of spectrum) during each pulse, 1/2 ≤ ((30 MHz) / (10 μ s)) MHz = 3 measurement bandwidth should be ≈ 1 a measurement bandwidth close to 1.73 MHz. In accordance with footnote hould be used in this example. but less than or equal to 1 MHz s the result of a measurement is as fo llows: for radars operating with multiple ≤ waveforms the measurement bandwidth is determined empirically fro m observations of the radar emission. Th d e empirical observation is performe as follows: the measurement system receiver is tuned to one of the fundamental frequencies of the radar, or is tuned to the centre frequency within the chirp range of the radar. The measurement system bandwidth is set to the widest available value, and the received power level from the rada r in this bandwidth is recorded. The measurement bandwidth is then rogressively narrowed, and the received power level is recorded as p a function of the bandwidth. The end result is a graph or table showing measurement system bandwidth. measured power as a function of will be the bandwidth where the The appropriate measurement bandwidth r l is observed. If a reduction in powe first reduction of the full power leve level is observed immediately, then the widest available measurement bandwidth should be used. 1 In all cases, if the above derived measurement bandw idth is greater than 1 MHz, then the corrections described in § 3.2 should be used.
6 4 Rec. ITU-R M.1177-4 measurement system bandwidth. Video bandwidth ≥ Detector p ositive peak. 3.1 Measurements within the out-of-band domain Within the out-of-band (OoB) domain, the lim its given in Recommend ation ITU-R SM.1541 are d an IF bandwidth leading to a a relative power measurement an defined in dBpp. This is en if the measurement measurement bandwidth less than the reference bandw idth should be used. Ev done, since both the peak no correction needs to be bandwidth is less than the reference bandwidth of the spectrum and the data points within the OoB domain are measured using the same measurement bandwidth B . m Measurements should generally be made using a bandwid th that is close to but less than the specified reference bandwidth. This approach will minimize the measurement time but it also causes some um. Thus in marginal situations, where measurement of the true broadening of the measured spectr close in spectrum shape may be important, it is r ecommended that the close-in region within the OoB T or 0.2/ t as appropriate. domain should be re-measured using a maximum bandwidth of 0.2/ Measurements within the spurious domain 3.2 3.2.1 Correction of the measurement within the spurious domain B Where the measurement bandwidth, , a correction B , differs from the reference bandwidth, ref m factor needs to be applied to the measurements conducted within the spurious domain to express the results in the reference bandwidth. Then the following correction factor should be applied: B = Spurious level (measured in Spurious level, B ) ) + 10 × log( B B / ref m m ref NOTE 1 – This correction factor should be used except where it is known that the spurious is not noise-like, B where a factor between 10 and 20 log( B / ) may apply and may be derived by measurements in more than m ref ) one bandwidth. In all cases the most precise result w B asurement bandwidth ( ill be obtained using a me m equal to the reference bandwidth. For radars B operating above 1 GHz the reference bandwidth ( ) is 1 MHz. ref 3.2.2 Correction of the measurement data to the peak envelope power n in RR Appendix 3 are defined in a reference Within the spurious domain, the limits give B , with respect to the peak envelope power (PEP). Data recorded in dBpp within the bandwidth, ref spurious domain must be refe renced to the PEP (and not the spectrum peak observed in dBpp). The PEP is approximated using the following correction formulae: For continuous wave (CW) and phase coded pulses: PEP = P + 20 × log( B B / B > ) for B pep pep m meas m For swept-frequency (FM or chirp) pulsed radars: 2 2 for )) /( log( 10 1 / < × × ( = ) B T B T B B P PEP + mes c m m c
7 Rec. ITU-R M.1177-4 5 where: PEP : peak envelope power; B : spectrum peak power ( P ); m meas : bandwidth calculated according to the following: B pep – for fixed-frequency, non-pulse-coded radar, one divided by the radar pulse radar pulse length is 1 μ s, then B μ is 1/1 length (s) (e.g. if the s = 1 MHz); pep – for fixed-frequency, phase coded pu lsed radar, one divided by the phase μ s long, then B e phase coded chip is 2 is chip length (s) (e.g. if th pep 1/2 μ s = 500 kHz); – for FM or chirped radar, the square root of the quantity obtained by dividing μ s) (e.g. if the FM is from the chirp bandwidth in MHz by the pulse length ( μ s, then B 1 250 MHz to 1 280 MHz or 30 MHz during the pulse of 10 is pep 1/2 = 1.73 MHz); s) μ (30 MHz/10 – for radars operating with multiple waveforms B is determined empirically pep from observations of the radar emi ssion. The empirical observation is performed as follows: the measurement system receiver is tuned to one of the fundamental frequencies of the radar, or is tuned to the centre frequency within the chirp range of the radar. The measurement system bandwidth is the received power level from the radar set to the widest available value, and e measurement bandwidth is then in this bandwidth is recorded. Th ived power level is recorded as progressively narrowed, and the rece result is a graph or table showing a function of the bandwidth. The end measured power as a function of measurement system bandwidth. bandwidth in which the full power The required bandwidth is the smallest B can be calculated from a knowledge of the level is still observed and pep impulse response of the measurement r eceiver using the criteria described below. If a reduction in power level is observed immediately, then the widest available bandwidth should be used. The corrections described in § 3.2 are illustrated gr aphically in Fig. 1. As can be seen in Fig. 1, the OoB mask and the measured spectrum have been referenced to the ). The Figure shows that the measured B / equivalent PEP level by using the factor 20 log( B pep m spurious is shifted upwards by an amount equal to the correction factor described in § 3.2.1 B (here taken as 10 log( / B )). In this example, a measuremen t bandwidth of 100 kHz was chosen ref m only for illustrative purposes, even though a bandwidth close to 1 MHz is recommended in this case. Also for illustrative purposes, the mask is shown offset in frequency as permitted in Recommendation ITU-R SM.1541.
8 6 Rec. ITU-R M.1177-4 FIGURE 1 Graphical illustration of the correction described in § 3.2 PEP 20 log( ) / B B pep m Example: = 20 MHz B pep OoB mask = 1 MHz B (Recommendation re f ITU-R SM.1541) B -40 = 100 kHz B m ) = 10 dB / 10 log ( B B m re f Spurious domain / B B ) = 46 dB 20 log ( e mission limit pep m (RR Appendix 3, Edition 2008) Attenuation of spurious emission power: PEP 43 + 10 log ( ), or 60 dBc, whichever is less stringent Spurious corrected Measured B / B ) + 10 log ( spectrum re f m OoB domain Spurious domain OoB domain Spurious domain 4 Measurements for multiple pulse or multimode radars dB bandwidth should be calculated for each For radars with multiple pulse waveforms, the B –40 individual pulse t ype and the maximum B dB bandwidth obtained shall be used to establish the 40 − shape of the emission mask (see Recommendation ITU-R SM.1541, Annex 8). For radars with multiple pulse width settings, that can be selected individually, the setting which dB bandwidth (see Recommendation ITU-R SM.1541, B results in the widest calculated − 40 Annex 8) should be used. Emission measurements onl y need to be carried out for this pulse width setting. For radars using elevation beam scanning, meas urements normally need only be made in the azimuth plane. Dynamic range of the measurement system 5 The measurement system should be able to meas ure levels of unwanted emissions as given in RR Appendix 3. To obtain a complete picture of the spectrum especially in the spurious emissions domain, it is recommended to be able to measure levels of emissions 10 dB below the levels given in RR Appendix 3.
9 Rec. ITU-R M.1177-4 7 e measurement dynamic range of the system should For a high level of confidence in the results, th be significantly higher than the required ra nge of measurement (margin (2) in Fig. 2). ecommended dynamic range of the The link between the required ra nge of measurement and the r measurement system is given in Fig. 2. FIGURE 2 Relation between the required range of measurement and the recommended dynamic range of the measurement system Measurement Levels to measure system dynamic range Radar frequency (2) B Spurious domain Spurious domain OoB OoB N (1) Spurious domain Spurious domain emission limit emission limit (RR Appendix 3) (RR Appendix 3) B –40 10 dB margin 10 dB margin (2) Boundary Boundary (1): Recommended range of measurement (2): Margin NOTE 1 – It should be noted that Recommendation ITU-R SM.329 recommends, under category B limits, more stringent limits than those given within RR Appendix 3 in some cases. This should be taken into account when evaluating the required range of measur ement and the recommended dynamic range of the measurement system. 6 Direct methods Two direct methods described below can be us ed to measure unwanted emissions (OoB and spurious) from radar systems. The first method is manually controlled and the second method is automatically controlled. These two methods have been used to measure the emission characteristics of radar systems operating at freq uencies up to 24 GHz, transmitter output powers of several megawatts, and e.i.r.p. levels in the giga watt range. Taking safety aspects into account, these methods may also be carried out in an anechoic chamber.
10 8 Rec. ITU-R M.1177-4 Measurement environment conditions 6.1 Regarding the measurement distance, either near field or far field measurements can be made. Variation of the peak received signal should be ma de less than 3 dB using the absorber when the receiving antenna is moved D /2 H horizontally or vertically away from the point where maximum λ : height of the transmitting point, signal is received ( : measurement distance, λ : transmitting H D wave length). Regarding the measurement site, it is preferable to locate the transmitting and receiving antennas in a fairly high position such as on towers. Note that the height should be de termined considering the r and measurement antennas, and no reflective objects should be vertical beam width of the rada between the antennas. 6.2 Measurement hardware and software required for the two direct methods are shown in Block diagram of the type of measurement system . The first element to be considered in the Fig. 3 (manual method) and Fig. 4 (automatic method) system is the receive antenna. The receive ante nna should have a broadband frequency response, at least as wide as the frequency range to be measured. A high-gain response (as achieved with so desirable. The high gain value permits greater dy namic range in a parabolic reflector) is usually al ovides discrimination agai the measurement; the narrow antenna beamwidth pr nst other signals in ems with multipath propagation from the radar the area; the narrow beamwidth minimizes probl under measurement; and spectrum da parabolic antenna require a minimum of ta collected with a post-measurement correction, as disc ussed in the next paragraph. Th e antenna feed polarization is selected to maximize response to the radar signal. Circular polari zation of the feed is a good choice for cases in which the rada r polarization is not known . The antenna polarization may be a priori on is used) or by excha tested by rotating the feed (if linear polarizati nging left and right-hand polarized feeds, if circular polarization is being used. Corrections for variable an tenna gain as a function of frequenc y should be consider ed. Antenna gain a theoretically perfect is levels are usually specified relative to that of otropic antenna (dBi). The effective aperture of an isotropic antenna decreases as 20 log( f ), where f is the frequency being measured. This means that, if the measurement antenna has a constant effective aperture (that is, has le antenna gain need be f )), no corrections for variab an isotropic gain that increases as 20 log( performed. This requirement is met by a theoretically perfect parabolic reflector antenna, and is one d for a broadband radar sp ectrum measurement. of the reasons that such an antenna is preferre Conversely, to the extent to wh ich the gain of the measurement antenna deviates from a 20 log( f ) curve (including a less-than-ideal parabolic antenna), the resulting m easurements must be corrected for such deviation. ement antenna to the measurement system should also be The cable connecting the measur considered. A length of low-loss RF cable (which will vary depending upon the circumstances of measurement system geometry at each measurem ent radar site) connects the antenna to the RF front-end of the measurement system. As losses in th is piece of line attenua te the received radar signal, it is desirable to make this line le ngth as short, and as low-loss, as possible.
11 Rec. ITU-R M.1177-4 9 FIGURE 3 Bloch diagram for measurement of radiated unwanted emissions from radars using the manually controlled direct method Ra nna d a r an t e e nt ant e nna: M e a s u r e m (rotating normally, or ( parabolic , w ith appropriate feed) sta tionar y and aligne d for maxi mum response in measurement equipment) Low -loss R F line (as short as possible bet ween antenna and me asu r em en t sy stem input port) M easurement s st em RF front-end y Optional notch, bandpass, or R1 Optional other filter or bandpass f ilter R2 Filter (used to attenuate r adar centre Optiona l V ariable R F atte nua tor frequency for measurements low-noise used to optimiz e at radar harmonic p r eamp li fi er measurement system frequencies) (LNA) gain/noise fi gure tr ade -off S elective R1 receiver Spectrum R2 analyzer
12 10 Rec. ITU-R M.1177-4 FIGURE 4 of radiated unwanted emissions Block diagram from measurement from radars using the automatically controlled direct method Measurement antenna: (parabolic, with appropriate feed) Radar antenna (rotating normally,or stationary and aligned for maximum response in measurement equipment) Noise diode calibration performed at this point Low-loss RF line (as short as possible between antenna and measurement system input port) Measurement system RF front-end Optional notch, bandpass or R other filter Filter (used to attenuate radar centre Low-noise Va r ia b le R F frequency for measurements preamplifier (LNA) attenuator at radar harmonic variable RF Tracking bandpass frequencies) (GPIB control filter from computer) (e.g., YIG filter) YIG tracking GPIB bus voltage GPIB bus R Spectrum PC-type computer analyzer LNA used to Fixed RF attenuation Control of improve spectrum used to optimize measurement system analyzer noise figure measurement system and recording of data gain/noise figure trade-off GPIB: general purpose interface bus YIG: yttrium-iron-garnet The RF front-end is one of the most critical pa ent system. It performs rts of the entire measurem three vital functions. The first is control and extension of measurement system dynamic range through the use of variable RF attenuation. Th e second is bandpass filtering (preselection) to prevent overload of amplifiers by high-amplitude signals that ar e not at the tune d frequency of the measurement system. The third is low-no ise preamplification to provide the maximum sensitivity to emissions that may be as much as ured level at the radar 130 dB below the peak meas fundamental. Each of these sections in the RF front-end is considered below. The RF attenuator is the first element in the front-end. It provides variable attenuation (e.g. 0-70 dB) in fixed increments (e.g. 10 dB/attenuat or step). Use of this attenuator during the measurement extends the dynamic range of the measurement system by the maximum amount of attenuation available (e.g. 70 dB for a 0-70 dB attenuator).
13 Rec. ITU-R M.1177-4 11 Manually controlled measurement system 6.2.1 The manually controlled measurement consists of sweeping across the spectrum in fixed increments (equal to the span value). At each frequency sweep, the attenuator is adjusted to keep the radar peak the other elements in the measurement system (often the power within the dynamic range of amplifier are the limiting elements). With the front-end amplifier and the spectrum analyzer log front-end RF attenuator properly adjusted at each sw eep, a measurement of the radar power at that frequency is performed. A manually controlled bandpass filter can be used at this point to avoid overloading the to measure very low spurious -compression), if it is necessary preamplifier (and thus causing gain emissions (i.e. level of fundamental emissions – level of spurious emissions > instantaneous measurement dynamic range). An LNA is installed as the next element in the The final element in the RF front-end is an LNA. e input characteristic of the LNA provides high signal path after the preselector. The low-nois , and its gain allows for the noise figure of the sensitivity to low-amplitude spurious radar emissions of transmission line and rest of the measurement system (e.g. a length a spectrum analyzer). The sensitivity and dynamic range of the measurement system are op timized by proper selection of LNA gain and noise figure character istics. It is desirable to mini mize noise figure while providing after the LNA (essentially enough gain to allow for all measurement circuitry the RF line loss after the front-end, plus the noise figure of the spectrum analyzer circuitry). Ideally, the sum of the LNA gain and noise figure (which is the ex cess noise produced by the LNA with a 50 Ω termination on its input) should be approximately equal to the noise figure of the remaining measurement system. Typical spectrum analyzer noise figures are 25 -45 dB (varying as a function of frequency), and transmission line losses may typically be 5- 10 dB, depending upon the qua lity and the length of the variation in measurement system noi tion of frequency, the line. As a result of se figure as a func a variety of LNAs used in frequency oc taves (e.g. 1-2 GHz, 2-4 GHz, 4-8 GHz, 8-18 GHz, 18-26 GHz and 26-40 GHz) may be required. Each LNA can be optimized for gain and noise figure within each frequency octave. This also helps match LNAs to octave breaks between various YIG LNA after the preselector (and, if required, filters (e.g. 0.5-2 GHz, 2-18 GHz, etc.). Use of an may reduce the overall measurement system noise a cascaded LNA at the spectrum analyzer input) figure to about 10-15 dB. This noise figure ra nge has been found to be adequate for the measurement of broadband radar emission sp ectra over a range as large as 130 dB. The remainder of the RF measurement system is expected to be esse ntially a commercially available spectrum analyzer or a spectrum analyzer with a preselector or a selective receiver. Any equipment, which can receive signals over th e frequency range of interest, can be used. 6.2.2 Automatically controlled measurement system ely in a radar measurement, as shown in Fig. 3, The key to using the RF front-end attenuator effectiv is to tune the measurement system in fixed-fre quency increments (e.g. 1 MHz), called steps, rather um, as is more conventionally done with manually controlled than to sweep across the spectr spectrum analyzers. At each fixed-frequency step, the attenuator is adjusted to keep the radar peak power within the dynamic range of the other elements in the measurement system (often the front-end amplifier and the spectrum analyzer log amplifier are the limiting elements). With the front-end RF attenuator properly adju sted at each step, a measuremen t of the radar power at that frequency is performed. In this way, a nominal 60 dB dynamic range for the measurement system is extended by as much as 70 dB, to a total resulting dynamic range of 130 dB. To minimize measurement time, this attenuator and the stepped-frequency measurement algorithm that it necessitates can be controlled by computer.
14 12 Rec. ITU-R M.1177-4 e tunable bandpass filter preselecto r is necessary if it is needed The next element in the front-end, th adjacent to much higher- at frequencies that are to measure low-power spurious emission levels level fundamental emissions (e.g. 130 dB below funda mental). For example, it may be necessary to measure spurious emissions from an air traffic control radar at 2 900 MHz that are at a level of undamental emission level is at +10 dBm and is –120 dBm in the measurement circuitry, while the f only 150 MHz away in frequency (at 2 750 MHz). The measurement system requires an unattenuated LNA to measure the spurious emission at 2 900 MHz, but the amplifier will be overloaded (and thus gain-compressed) if it is exposed to the unattenuated funda mental emission at 2 750 MHz. For this reason, attenuation that has fr equency-dependence is required in the front-end at a position before the LNA input. In practice, this tunable bandpass filtering is effectively provided by varactor technol G technology (above 500 MHz). ogy (below 500 MHz) and by YI ly, and should be designed to automatically track The applicable filters may be procured commercial the tuned frequency of the measurement system. The final element in the RF front-end is an LNA. An LNA is installed as the next element in the signal path after the preselector. The low-nois e input characteristic of the LNA provides high sensitivity to low-amplitude spurious radar emissions , and its gain allows for the noise figure of the of transmission line and a spectrum analyzer). rest of the measurement system (e.g. a length ity and dynamic range of the measurement system, as well as for Considerations for the sensitiv es, are the same as stated in § 6.2.1. typical spectrum analyzer noise figur Another option for LNA configuratio n is one in which LNAs are cascaded. The first LNA is placed ndpass preselector filter. It has a low noise figure, between two stages within the YIG or varactor ba but only enough gain to allow for the insertion loss of the sec ond YIG stage. A second (possibly lower-performance) LNA is placed immediately afte r the YIG. This option will provide somewhat lower overall system noise figure because the sec ond stage of the YIG is allowed for by the first ced design and engineering modifications to the LNA. However, this option may require more advan preselector filter than an ad ministration may deem practical. requiring any redesign onfiguration, and one not A third option for the measurement system LNA c or retrofitting of the front-end preselector filter , is to place a lower-gain LNA in the front-end and a second LNA at the spectrum analyzer signal input. The first LNA is select ed to have very low to allow for the RF lin noise figure and just enough gain re of the spectrum e loss and the noise figu analyzer LNA. The spectrum analyzer LNA, in turn, aracteristic that is just is selected for a gain ch analyzer’s noise figure in the a adequate to allow for the spectrum ppropriate frequency range of the radar measurement. This set of two cascaded L NAs may be more easily acquired than a single, extremely high-performance LNA, and will typically be less susceptible to overload as the 1 dB compression points can be expected to be higher than those for individual high-performance LNAs. The remainder of the RF measurement system is expected to be esse ntially a commercially which can receive signals over the frequency available spectrum analyzer. Any spectrum analyzer -controlled to perform the stepped-frequency range of interest, and which can be computer noise figure of currently algorithm, can be used. As noted above, the high available spectrum analyzers must be allowed for by low-noise preamplification if the measurement is to achieve the necessary sensitivity to observe most spurious emissions. The measurement system can be controlled via any computer which has a bus interface (GPIB or equivalent) that is compatible with the comput er controller and interface card(s) being used. In terms of memory and speed, modern PC-type computers are quite adequate. The measurement algorithm (providing for frequency stepping of the spectrum analyzer and the preselector, and control of the front-end variable attenu ator) must be implemented through software. Some commercially available software may approach fulfilment of this need, but it is likely that the measurement organization will need to write at l east a portion of their own measurement software. While the development of software requires a si gnificant resource expenditure, practical experience
15 Rec. ITU-R M.1177-4 13 to be worthwhile if radar emission measurements with such systems has shown such an investment are to be performed on a fre quent and repeatable basis. eally, a data record is e or on a removable disk. Id Data may be recorded on the computer’s hard driv data files manageable, and to eps, so as to keep the size of made for every 100-200 measurement st prevent the loss of an excessive amount of data if the measurement system computer or other components should fail during the measurement. 6.3 Measurement system calibration Manual direct method 6.3.1 The manually controlled method re quires either calibra asuring components tion of all the me ith a calibrated generator (substitution method). individually or of the whole measuring set w Automatic direct method 6.3.2 The measurement system is calibrated by disconnectin g the antenna from the rest of the system, and point. A 25 dB excess noi attaching a noise diode to the RF line at that se ratio (ENR) (where ENR = (effective temperature (K), of noise diode/a mbient temperature (K)) diode should be more than adequate to perform a satisf actory calibration, assuming that the overall system noise figure is less than 20 dB. The techni que is standard factor, Y , measurement, as descri bed in Appendix 2 to Annex 1, with comparative power measurements made across the spectrum , once with the noise diode on and once with the noise diode off. The noise diode calibration results in a table of noise figure values and gain corrections for the entire spectral range to be measured. The gain corr ections may be stored in a look-up table, and are applied to measured data as those data are collected. Appendix 2 to Annex 1 describes the calibration procedure in more detail. in the field. Correction factors for the antenna The measurement antenna is not normally calibrated (if any) are applied in post-measurement analysis. 6.4 Measurement procedure 6.4.1 Manual method Appendix 1 to Annex 1 describes the direct method in detail; this section pr ovides a summary of the method. Prior to measurement, a spectrum analyzer is used to detect the presence of signals not emitted by the radar: if there are emissions corrupting the measurement, appropriate filters must be used. Max-Hold function Spectrum analyzer lowest frequency to be measured (e.g. if radar centre frequency is centre frequency 3 050 MHz, but the spectrum is to be measured across 2-6 GHz, then initial spectrum analyzer centre frequency would be 2 GHz). Spectrum analyzer 10, 20, 50, 100, or 500 MHz. = frequency span Spectrum analyzer automatic sweep time > sweep time
16 14 Rec. ITU-R M.1177-4 3 radar beam rotation intervals. record signal during a minimum of Time > (e.g. if radar rotates at 40 r.p.m. or 1.5 s per rotation, then duration shoul d 3 × 1.5 s; 4.5 s would be a reasonable selectio n). Record signal fo be > r . Radar antenna may a sufficient time for spectrum to form be held stationary and aligned for the maximum measurement system response. time and the signal record duration should be NOTE 1 – The setting of the spectrum analyzer sweep validated. The second measurement point is taken by tuning the measurement system to the next frequency equal to the first measured frequency band plus band to be measured. This frequency is optimally the measured span. In the case where the measurement instrument is a selective receiver, the m easurement is done point by point according to the recommended bandwidth. 6.4.2 Automatic method Appendix 1 to Annex 1 describes the direct method in detail; this section provides a summary of the § 2, the spectrum analyzer should be set up as method. In addition to the parameters listed in follows: lowest frequency to be measured (e.g. if radar centre frequency is Spectrum analyzer 3 050 MHz, but the spectrum is to be centre frequency measured across 2-6 GHz, then initial spectrum analyzer centre frequency would be 2 GHz). Spectrum analyzer a time-domain instrument). 0 Hz (analyzer is operated as = frequency span Spectrum analyzer radar beam rotation interval (e.g. if radar antenna rotates at 40 r.p.m. o r > step time 1.5 s/rotation, then step time should be 1.5 s; 2 s would be a reasonable > or radars with vertical scanning selection). For frequency agile radars ve to be several antenna rotatio n antenna beams, the step time may ha p eriods. For these more complex radar systems, the step time should be determined empirically. and with the measurement system set up as With the radar antenna beam scanning normally, described above, the first data point is collected. A data point consists of a pair of numbers: h the power level was measured. For example, measured power level and the frequency at whic 000 MHz. The data point is the first data point for the above measurement might be − 93 dBm at 2 collected by monitoring the radar emission at the desired frequency, in a frequency span of 0 Hz, for an interval (step time) slightly longer than that of the radar antenna rotation period, or for a longer step time for complex radar systems. This time-display of the radar antenna beam rotation een. The highest point on the trace will normally will be displayed on the spectrum analyzer scr was aimed in the direc tion of the measurement represent the received power when the radar beam system. That maximum received power value is retrieved (usually by the control computer, although it could be written down manually), corre cted for measurement system gain at that frequency, and recorded (usually in a data file on magnetic disk). The second measurement point is taken by tuning the measurement system to the next frequency to be measured. This frequency is optimally equa l to the first measured frequency plus the measurement bandwidth (e.g. if the first meas urement was at 2 000 MHz and the measurement bandwidth were 1 MHz, then the second measured frequency would be 2 001 MHz). At this second frequency, the procedure is repeated: measure th e maximum power received during the radar beam rotation interval, correct the value for gain f actor(s), and record the resulting data point.
17 Rec. ITU-R M.1177-4 15 her than sweeping) across the spectrum, continues This procedure, which consists of stepping (rat e stepping proce ss consists of until all of the desired emissi on spectrum has been measured. Th measurements made at predetermined (fixed-tuned) frequencies a series of individual amplitude across a spectrum band of interest. The frequency change between steps is optimally equal to the measurement system IF bandwidth. For example, measurements across 200 MHz of spectrum might use 200 steps at a 1 MHz step interval and a 1 MH z IF bandwidth. The step interval may be set wider in the spurious emissi on domain to expedite the ove rall measurement. However, .g. 2, 3, 4) of the fundamental radar emission, at frequencies that are integral multiples (e the maximum step interval should again be about equal to the measuremen t system IF bandwidth. equency for a specified measurement interval. The measurement system remains tuned to each fr dwell for each step is specified by the measurement The interval is called step time, or dwell. The system operator, and is normally slightly l onger than the radar beam scanning interval. sirable if this process (step, tune, measure, Computer control of the measurement system is de be performed with efficiency a nd accuracy. In order to correctly correct for gain, and repeat) is to measure the peak of the fundamental emission it may be required to use a smaller step interval of the order of half or less of the m easurement bandwidth over this region. The stepped time technique is requi red to enable the insertion of RF attenuation at the front-end of the measurement system as the frequencies appro ach the centre frequency (and any other peaks) of on on a frequency-selective basis makes it possible the radar spectrum. This ability to add attenuati to extend the dynamic range available for the measurement to as much as about 130 dB, if a 0-70 dB RF attenuator is used with a measurement system having 60 dB of instantaneous dynamic range. This is of great benefit in iden tifying relatively low-power spurious emissions. To achieve the same effect with a swept-frequency measurement, a notch filter could be inserted at the centre frequency of the radar, but there would be no practical way to insert a notch filter for all the other high-amplitude peaks that might occur in the spectrum. It is important to provide adequate bandpass filt ering at the front-end of the measurement system, so that strong off-frequency signal component s do not affect the meas urement of low-power spurious components. These measurements may be performed without the radar beam being scanned in space, but only if it is verified that the direction of the radar beam relative to the mechanical axis of the antenna does not vary across the frequency range of the measurement. 6.4.3 Indirect method component separation for the Indi rect method. In this Indirect Figure 5 illustrates a recommended method, where unwanted emissions ar t and then, combined with the e measured at the rotating-join antenna characteristics measured separately at di stances of 5 m and 30 m w ith appropriate far-field correction, the procedure is: Step 1 : Make measurements of a radar transmitter emissions at the Ro-Jo with a feeder (as shown in Fig. 6). Step 2 : Then make separate measurements of a radar antenna maximum gain at the emission frequencies found in Step 1. Here, measurements ar e made at the distances of 5 m for frequencies below 5 GHz and 30 m for frequencies above 5 GHz (as shown in Fig. 7).
18 16 Rec. ITU-R M.1177-4 FIGURE 5 Typical radar system Antenna Coaxial cable Recommended component separation for indirect method Radar Radar transmitter tr an s m itt e r Rotating joint Transmitter output in WG 10 WG: waveguide FIGURE 6 Measurement at the Ro-Jo part EIA right-angle Ro-Jo adaptor Short or long feeder Waveguide to Radars EIA adaptor Special attenuator Waveguide transitions WG 10-12 WG 10-14 WG 10-16 WG 10-18 Measuring cable Waveguide to Spectrum -type N analyzer adaptors A coaxial attenuator or a notch filter is needed in WG 10 and WG 12 to further enhance the measuring sensitivity EIA: Electronics Industry Associations
19 Rec. ITU-R M.1177-4 17 : Correct the measured gains with an appropria te correction factor (u Step 3 sing a software program e simplest cases it may be possible to use the or model of the known antenna performance. In th software programme, given in Appendix 4 to Anne x 1, for the frequencies, at which the emissions were observed in Step 1). the effective e.i.r.p. ra diation at the observed Step 4 : Finally, Steps 1 and 3 are combined to obtain unwanted emission frequencies. Methods of measurement and proble ms associated with a waveguide 126.96.36.199 There are two main problems in measuring the transmitter output power spectrum. The one is accessing the higher frequency components of the transmitted spectrum without distortion and; in the presence of the fundamental transmitting the other is measuring very low level emissions pulse of perhaps 60 kW peak power. In any waveguide, the propagation mode, TE , can be measured using a calibrated measuring 10 s the powerful fundamental system. The characteristic of such a system must be such that it attenuate quipment, whilst at other frequencies offers signal sufficiently to protect the measuring e gy is being measured in the TE a negligible attenuation and ener mode. 10 It should be recognized that, the spurious frequenc y emissions of the transmitter output could be in higher order modes and this possi bility should be considered when setting up the measurement system. For simple radars however this will rarely be of significance as such higher order modes are generally trapped in a wa veguide to coaxial adapto r, or in antenna feeder and the Ro-Jo connecting to the radar antenna. (i.e., wavegui de to coaxial adaptors are onl y designed to couple energy in TE mode). 10 urement of unwanted emissions in a 188.8.131.52 The measurement system for the meas waveguide This measuring system allows the accurate measurem ent of low levels of emissions in the presence of high power radar pulses. The main components of the system are a notch filter a nd a set of waveguide tapers, from WG 10 to smaller waveguide sizes, to cover the whole frequency range of interest. The notch filter comprises ements inside, which attenuates the fundamental of a straight WG 10 waveguide with absorbent el offers negligible attenuation. To signal while at other frequencies it ired attenuation achieve the requ e emissions at higher fr to protect the measuring equipment, and to measur equencies, linear tapers are used at the output of the notch filter. The waveguide taper is a high pass filter and thus re jects, by reflecting back, signals below the cut off frequency. If a taper had been used directly at an output port of a radar transmitter, the fundamental would have been reflected b ack into the transmitter causing an undesirable the notch filter the reflected si gnals are absorbed a second time. mismatch. But with the taper after Thus the return loss at the fundamental frequenc h is low enough to avoid y is typically 34 dB, whic frequency pulling of the magnetron. Frequencies above the cut off are transmitted th rough the transitions and into the measuring equipment. If possible, a short waveguide sec tion, should be included to prevent coupling of evanescent modes between a taper and axial transition. a waveguide to co 184.108.40.206 Results of measurement at the Ro-Jo port The measurement technique comprises an explorator y search of a frequency band of interest to locate and tag significant spurious emissions by frequency, followed by a revisit to each noted emission for detailed and accurate measuremen t of maximum amplitude of that emission.
20 18 Rec. ITU-R M.1177-4 Measurement uncertainty in a waveguide 220.127.116.11 1.3 dB across the frequency band 2 to 18.4 GHz for The system has a measurement accuracy of ± the waveguide port. Total uncertainties with a confidence level of not less than 95% can be ± cluding the spectrum analyzer. calculated to be 3.4 dB for the waveguide port in 18.104.22.168 Measurement of antenna gain characterist ic at measured emission frequencies This indirect method recommends that near-field measurements be made on the antenna on an open ce of 5 m for frequency below 5 GHz area test site (OATS) at distan and 30 m for frequencies above 5 GHz. Correction factors are then applied to correct the measuremen t to an equivalent far field gain, which provide an acceptable correlation w ith the far field gain. A typical measurement arrangement is shown in Fig. 7. FIGURE 7 ement for 5 m and 30 m distances Near field gain measurement arrang Separation distance: 5 m for frequencies less than 5 GHz 30 m for frequencies above 5 GHz Height search Radar antenna TX cable 1-4 m Calibrated test horns Directional coupler RX cable Fixed height Signal 1.5 m Measuring generator equipment Turntable Earth plane Antenna mast 22.214.171.124 Near field gain measurement proc edure for 5 m and 30 m distances The measurement of maximum gain of the antenna unde r test (AUT) shall be carried out at spurious and OoB frequencies measured or identified, using the met hod specified in § 6.4.3. At each in of AUT shall be maximized by first rotating measured, or identified, emission frequency, the ga ng the test horn up, or down. The gain of the through 360° and then further maximized by movi AUT is obtained by measuring e.i.r.p. at each dist ance with a known level of power into the AUT ) show details of calculations to arrive at the at each frequency of interest. Equations (1) and (2 , of the AUT from the measured spectrum analyzer level, S . G equivalent far field gain, a G (1) of the AUT (dBi) = measured e.i.r.p. (dBm) – P (dB) (dBm) + G c a input d 4 π + (2) (dBi) G (dBm) (dB) – Measured e.i.r.p. (dBm) S 20 log = r λ
21 Rec. ITU-R M.1177-4 19 where: : equivalent far field gain of the AUT (dBi) G a P : power input into the AUT (dB) input distances, which can be calculated for : gain correction factors for 5 m and 30 m G c ecified in Appendix 4 to Annex 1 the AUT using a software program sp : measured spectrum analyzer level (dBm) S G : gain of the receiving test horn antenna (dBi) r d : measuring distance (m) λ : wavelength of a frequency of interest (m). Gain correction and reduction factors 126.96.36.199 nnex 1 gives the far field correction factors from The software program given in Appendix 4 to A a near field measurement for a very simple case. The program derives the co rrection factor for each distance at the frequency of interest by consideri ng the phase changes of the received wave across is spherical and not linea r.) Therefore, it can be the linear antenna. (At near distances the wave front used to infer the maximum antenna gain at infinity from a near field measurement. An important point to bear in mind not addressed. It must be noted is that the antenna gain pattern is that at spurious frequencies the electrical length of the antenna is different from the mechanical length; it may well be much shorter. This is due to the different illumination pattern of the antenna length at frequencies other than the designed frequency. Thus a mo re complex software model or data derived using the direct method may be require d to achieve accurate results in such cases. 188.8.131.52 Near field gain measurement uncertainty with the applied correction factors inty can be calculated to be The worst-case measurement uncerta ± 6 dB, which includes, uncertainties due to a spectrum analyzer, test hor n gain, cable loss and sour ce and site imperfection. not less than 95% can be calculated to be ± Total uncertainties with a confidence level of 4.2 dB. The derivation of the correction f actors for these distances assumes the AUT radiating aperture to be constant at all frequencies. 184.108.40.206 Producing a radar transmitter emission spectrum as an e.i.r.p. by combining measured emissions and antenna gain characteristic r omnidirectional e.i.r .p. is to add, for each The technique used to obtain a maximum value fo emission frequency, the maximum power generated by a radar transmitter (dBm), to the maximum directional gain (dBi) from the AUT. This means one only has to characterize the AUT at frequencies at which the radar transmitter emissions were observed. The effects of the AUT mismatch account automatically in the are considered to be taken into Ω , the nominal impedance of measurements of gain, because the test equipment is matched to 50 issions are measured in the 50 Ω measuring receiver. the coaxial connectors and the em 220.127.116.11 Summary The indirect method, which is cost effective in time and facilities, is sensitive enough to allow measurement of low level emission values with a reasonable accuracy and repeatability. Furthermore, it can be used in all weather co nditions and the measurement frequency range can easily be extended to 40 GHz or higher. It can also usefully be used in c onjunction with the direct method to assess incremental changes in a given ra dar system that has been previously measured.
22 20 Rec. ITU-R M.1177-4 Appendix 1 to Annex 1 Direct method detailed description of procedures and software The direct method assumes that the following conditions can be met: – the far field radiation zone of a radar can be accessed by a measurement system as described in the body of this Annex; r signals directly into the measurement system hardware – unwanted feed-through of rada (i.e. bypassing the measurement system antenna) can be minimized to a sufficiently low level to ensure that measurement results are accurate. r operation be coordinated with the measurement The direct method does not require that the rada system, although in some cases cooperative opera tion may be beneficial in expediting the measurement. The direct method process is as follows: Step 1: determine a measurement location The measurement location should be within or as n ear as possible to the ra dar main radiation beam. For surface search radars and some other radar types, this may be relatively easy, as the radar beam will sweep across the surface, and the measurement sy stem need only be placed within this area. For many air search radars, howev er, the main beam does not dir ectly illuminate the ground. For these radars, the measurement system should be located within the maximum coupling zone on the surface. This zone may be determined by tuning the measurement system to the radar fundamental frequency and then driving the measur ement system in a vehicle from a position close to the radar to a position far ( on the order of a few kilometres) from the radar. The measurement l as a function of positi system is used to monitor received signal leve on. This can be done by running a spectrum analyzer in a zero frequency sp an with a sweep time of 500 s, and watching the peak level every few seconds when the radar sweeps past the vehicle. The re sult is a time display that shows the maximum coupling location(s). adequate. In practice, this zone has been Any place within the maximum coupling zone should be found to begin no closer than about 0.75 km from ai r search radars, and to extend to no further than about 2 km from the same radars. There is usually no sharply define d point where maximum coupling occurs, but rather a br oad zone within these limits. The question of multipath should be considered. Mult observed very rarely. ipath effects have been When they have been observed, it has been in ca ses in which the radar a nd the measurement system were separated by calm, smooth wate r surfaces. In other cases, irre gular intervening terrain and the use of parabolic reflector antennas by the meas urement system minimize multipath effects to an extent that makes them negligible. Multipat h effects can be checked by repeating the radar measurement at a second location and comparing the results from the two measurement locations. Multipath is also believed to be minimized by raising the measurement antenna on a telescoping mast to a height of about 10 m above the ground. This also provides a bette r line-of-sight between the radar and the measurement system.
23 Rec. ITU-R M.1177-4 21 and check for unwanted feed-through signals Step 2: set up the measurement system the top of a 10 m mast The measurement system is configured with a para bolic reflector antenna at tres above the ground, to avoid multipath effects and (optional), or at a height of at least a few me provide reasonably good lin e measurement system should be tuned to the e-of-sight propagation. Th radar fundamental frequency or maximum emissi on frequency, if it is chirped or frequency- hopping. It is necessary to check for unwanted feed-thr ough (i.e. the unwanted reception of radar energy within the measurement equipment, bypassing the measurement antenna). Feed-through is checked by disconnecting the measurement antenna and terminating the input line with a 50 Ω load. If feed- through is present, the following options may be exercised: check to ensure measurement equipment racks (if any) are sealed; – check connectors for firm fittings; – move the radar measurement system to an – alternative location, in which the measurement equipment is shielded from the radar by buildin gs or foliage, and in which the antenna is raised above these obstacl es on the telescoping mast; move the radar measurement system to a larger distance from the radar. – A well-designed measurement system should mini mize the possibility of unwanted feed-through. emissions parameters Step 3: determine radar The parameters that are most critical to dete rmine before the measurement begins are beam ndwidth. Beam scanning interval and other scanning interval and effective emission ba characteristics are acquired by tuning the spectrum analyzer in a zero span mode and a sweep time interval of several seconds, and then observing the beam scanning of the radar. Determination of the emission bandwidth is acco mplished as described in the main body of this dar fundamental frequency in a zero span mode, Annex, with the spectrum analyzer tuned to the ra eir widest available valu es. The IF bandwidth is and the IF and video bandwidths initially set to th then reduced each time the radar beam swings pa st the measurement system, and the bandwidth at which the received power level drops is noted. This is the widest available measurement bandwidth will be the measurement bandwidth used, unless that is less than the radar emission bandwidth. This dar in a particular receiver bandwidth dictate circumstances such as a need to observe the ra otherwise. Additional radar emission parameters that should be noted are: pulse repetiti on rate, pulse jitter (if any), pulse stagger (if any) , and pulse width. The first thr ee of these parameters may be measured on an oscilloscope connected to the spectrum analyzer’s vide o output. The RF pulse width (50% voltage points) and ri se time (10-90% voltage points) should be measured with a peak operated in the square-law region. It should be power meter or suitable wideband RF detector diode, properly matched to an oscillosc enable display of the pulse ope having a sufficient bandwidth to waveform without distortion associated with limited detector bandwidth. Step 4: calibrate the measurement system Manually controlled direct method: – libration of all the measuring components The manually controlled method requires ca individually or cal ibration of the whol e measuring set. Automatically controlled direct method: – See Appendix 2 of Annex 1. Noise diode ca libration is recommende d, although alternative methods using signal generators can be used.
24 22 Rec. ITU-R M.1177-4 Step 5: configure measurement system software (automatic method only) The measurement software must be configured to the desired start frequency (MHz), stop frequency (MHz), step size (MHz), step interval (M Hz), IF bandwidth (MHz), video bandwidth ( IF bandwidth), detector (positive peak), spectr um analyzer reference level (usually –10 dBm), ≥ initial attenuation at the start frequency (usually 0 dB), and additional data regarding the location (such as radar name, project name for the measurement, etc.). Step 6: check for linear ity during the measurement the measurement by checking for linearity as the It is critical to maintain the integrity of measurement progresses. When measuring, both at the fundamental frequenc y and in the spurious odically inserting 10 dB emissions, system linearity should be checked by peri of RF attenuation at the RF front-end, ahead of the LNA. The result shoul d always be a 10 dB drop in measured signal either front-end overload level. If other than a 10 dB drop is observed or unwanted feed-through may be occurring. Good system design will minimi ze these potential problems. If they do occur, additional steps to shield the measurement system, it may alternatively be necessary to either take tion, as described in Step 2, above. or else to move to another loca Step 7: measure the radar in more than one IF bandwidth (recommended but not required) It may be useful to measure radar emissions in several bandwidths. Such measurements provide an unequivocal indication of the variation in m easured radar power as a function of receiver bandwidth at any given frequency in the spectrum. Appendix 2 to Annex 1 Gain and noise figure calibra tion using a noise diode Measurement system calibration should be perfor med prior to every radar emission spectrum measurement. As measurements are performed, ga in corrections may be added automatically to every data point. For measurement system noise figures of 20 dB or le ss, noise diode factor, Y , calibration (as described below) may be used. This Appendix describes the theory and procedure for such calibration. a particular frequency may be represented in The noise diode calibration of a receiver tuned to lumped-component terms as shown in Fi g. 8. In this diagram, the symbol Σ represents a power- summing function that linearly adds any power at the measurement system input to the inherent noise power of the system. The symbol g represen ts the total gain of the measurement system. The measurement system noise factor is denoted by nf, and the noise diode ha s an excess noise ratio denoted as enr. (In this Appendix, all algebraic qua ntities denoted by lower-case letters, such as “g”, represent linear units. All algebr aic quantities denoted by upper case le tters, such as “G”, represent decibel units.)
25 Rec. ITU-R M.1177-4 23 FIGURE 8 Lumped component diagram of noise diode calibration Noise Power diode summation System enr Antenna output g input Measurement system nf gain Measurement system inherent noise n (W), and thermal noise: noise power from a device, Noise factor is the ratio of device n device k T B where: − 23 × 10 J/K) k : Boltzmann’s constant (1.38 T : system temperature (K) : bandwidth (Hz). B the noise factor minus one, ma king it the fracti on of power in The excess noise ratio is equal to . The noise figure of a system is defined as 10 log (noise factor). As many noise excess of k T B sources are specified in terms of excess noise ratio, that quantity may be used. In noise diode calibration, the primary concern is the difference in output signal when the noise (W), is given by: diode is switched on and off. For th e noise diode = on condition, the power, P on × B k g enr nf p ( T + = ) on s d where: nf : system noise factor s enr : noise diode enr. d P When the noise diode is off, the power, (W), is given by: off T k g nf B ( ) × = p off s P The ratio between and P factor: is the Y off on + p ( ) nf enr on s d y = = p nf s off p on log 10 = 10 = log( ) P P Y − = y on off p off
26 24 Rec. ITU-R M.1177-4 Hence the measurement system noise factor can be solved as: enr d nf = s y 1 − The measurement system noise figure is: enr 10 / Y d = 10 NF log 10 y ) − − = − − = 1 10 ENR ( log 10 ENR ) 1 ( log d d s − y 1 Hence: − p p off on g = × B T enr k d ) ( log 10 ) ( log 10 B T k enr p p G = − − × on d off or P 10 / P 10 / off on = − 10 T k ENR G 10 log ( log 10 − − 10 B ) d used to calculate measurement system gain In noise diode calibrations, the preceding equation is from measured noise diode values. Although the equation for NF may be used to calculate the measurement system noise figure, s software may implement an equivalent equation: p off = nf s g B k T = ) ( log 10 ) ( log 10 ) ( log 10 B T k G P B T k g p NF − = − − s off off and substituting the expression for gain into the preceding equation yields: P / 10 P 10 / off on − + − = 10 log 10 NF P ENR 10 off s d The gain and noise figure values determined with these equations may be stored in look-up tables. The gain values are used to correct the measured data points on a freque ncy-by-frequency basis. is calibrated with a noise diode source prior to Excluding the receive antenna, the entire signal path ise diode is connected to the input of the first RF line in place a radar spectrum measurement. A no of the receiving antenna. The connection may be acco mplished manually or via an automated relay, io. The noise level in the system depending upon the measurement scenar is measured at a series of points across the frequency range of the system with the noise diode turned on. The noise measurement is accomplished with the IF bandwidth set to 1 MHz and the video bandwidth set to 1 kHz. The noise diode is then tu rned off and the system noise is measured as before, at the same frequencies. The measurement system computer thus collects a set of P and P values at a series on off of frequencies across the band to be measured. The values of P and P are used to solve for the on off gain and noise figure of the measurement system in the equations above.
27 Rec. ITU-R M.1177-4 25 Appendix 3 to Annex 1 and pulse rise/fall times Measurement of pulse width 1 Introduction This Appendix is intended to provide guidance fo r measuring radar pulse parameters needed in applying the emission mask fo r the OoB domain. Recommenda tion ITU-R SM.1541, Annex 8, for radar systems. To determine the necessary addresses unwanted emissions in the OoB domain bandwidth, , and the 40 dB bandwidth, B B , the pulse width, t , and the rise time, t , of pulsed r –40 n 2 radars must be measured . Pulse width, , is measured at the –6 dB points (50% voltage points) of a radar pu lse. The rise time, t , or fall time, , is measured between th e –0.9 dB and –20 dB (10%-90% voltage points) on t t r f a pulse’s leading or trailing edge , respectively. For coded pulses, t and t are the rise/fall times of r f are not discernable, then it may be assumed that a sub-pulse. If sub-pulses t is 40% of the time to r switch from one phase or sub-phase to the next. For some radar designs, pulse widt easured via a hardline connection h and rise or fall time may be m to a directional coupler. However, radiated pulse characteristics may differ somewhat from those measured from directional couplers. Moreover, some radar designs do not provide a directional coupler. For these radars, pulse widt h and rise or fall time can be m easured via radiated energy if t bandwidth (that is, exceeding (10/ the measurement system has sufficien ) or a bandwidth that can t r ine the true rise time). A potentia l impediment to measuring pulse be adequately corrected to determ width via radiation is the effect of multipath ener gy, which causes a stair-step fall-off on the trailing edge of each radiated pulse. This effect can be minimized by the use of a parabolic reflector antenna on the measurement system. If the effect of multip ath can be suppressed enough for the first trailing e nominal pulse level, then a radiated measurement edge stair step to occur more than 6 dB below th 3 e bandwidth requirement is met of pulse width is possible if th . A broadband diode detector is required to achieve sufficient bandwidth. Measurements for conventional radars 2 Hardline-coupled pulse measurements 2.1 teristics, the measurement setup is shown in For hardline-coupled measurements of pulse charac Fig. 9. A coaxial cable of appropriate impedan ce is connected between the directional coupler output and the input of a wideband (bandwidth exceeding (1/ t )) crystal detector. A variable r attenuator (0-70 dB, for example) is inserted between the coupler and the detector. Prior to connecting the detector, the attenuator is initially set to a sufficiently high level as to protect the 2 When the fall time, t , of the radar pulse is less than the rise time, t , it should be used in place of the rise r f time in applying equations in Recommendation ITU-R SM.1541. 3 For example, a 1 μs pulse might have a rise time of less than 0.1 μs. This t would require bandwidth in r excess of 10 MHz for accurate measurement. Oscilloscop es are available with bandwidths of up to 2 GHz. For purposes of measuring radar rise/fall times, oscill oscopes with at least 500 MHz should be used. The bandwidth must be available in a single-shot (not repetitively sampled) mode, as the measurements are made on single radar pulses.
28 26 Rec. ITU-R M.1177-4 4 . The maximum permissible detector input level may be assumed to be crystal from damage +20 dBm, if other data are lacking. FIGURE 9 Block diagram schematic for me asuring radar pulse width and rise time (or fall time) parameters via a hardline connection to a directional coupler Radar directional coupler High power output Attenuated output Attenuation as required to Calibrated signal generator that protect detector from burnout can approximately replicate and to obtain square law input radar pulse characteristics to detector Wideband oscilloscope Matched (fast enough to load T measure (1/ )) r cilloscope having bandwidth that exceeds (1/ The detector output is connected to an os t ). r Impedances should be matched appropriately; most modern oscilloscopes having selectable input impedance values. 50 is typically correct. DC coupling should be used on the oscilloscope input. Ω 5 The oscilloscope is adjusted to display and record radar pulse envelopes. The setting of the variable attenuato r is noted by measurement personnel. 4 The initial attenuator setting may be derived from the radar’s peak power level and the specified insertion loss of the directional coupler. 5 Most oscilloscopes can record data to either an inte rnal disk or to an external computer via an IEEE-488 (GPIB) bus. Recording may also be achieved by p hotographing the oscilloscope screen with a digital still-frame camera.
29 Rec. ITU-R M.1177-4 27 that device. It is reconnected Next, the line from the radar direct ional coupler is disconnected from ses of approximately l generator that is capable of producing pul to the output of a calibrated signa the same width as measured from the radar. The signal generator output is adjusted to generate the same for both envelopes, preferably about an amplitude response on the oscilloscope that is +10 dBm. With this adjustment made, the response of the crystal detector can be calibrated as follows. 6 The signal generator output is decreased by, successively, 0.9 dB, 6 dB and 20 dB . At each of measured pulse envelope. The resulting time these levels, vertical markers are placed on the rkers provide the pulse width ( Δ between the 6 dB points), the rise intervals between the vertical ma Δ between the 0.9 dB and 20 dB points on the leading edge), and the fall time ( Δ between the time ( on the trailing edge). 0.9 dB and 20 dB points Radiated coupled pulse measurements 2.2 For radars that do not incorporate directional coupl ers, pulse characteristics can only be measured through radiated measurements. Figure 10 shows the measurement hardware configuration for measuring radiated pulses. FIGURE 10 dth and rise time (or fall time if shorter) Block diagram schematic for measuring radar pulse wi parameters via radiated pulses Radar Optional Wideband Wideband transmitter attenuation or bandpass oscilloscope Matched amplification to filter (fast enough to load obtain square-law (wider than measure (1/ )) T r input to detector (1/ )) T r The procedure that should be used is as follows: Step 1: on with clear line-of-si ght path to the radar Position the measurement system at a locati transmitter antenna, and as close as possible wit hout suffering degradation to the measurement system performance (e.g. feed-thr ough), losing power from the radar by passing underneath the main beam, or being within the ne radar antenna or the measurement ar-field distance of either the antenna. Step 2: Use a high-gain antenna (e.g. a 1 m diameter or larger parabolic at microwave frequencies) on the measurement system to receive pulses from th e radar at the highest possible amplitude and to discriminate against signals from other transmitters. 6 Crystal detector outputs are not necessarily linear; therefore the 10%, 50% and 90% voltage points for the RF signal may not appear to be 10%, 50%, and 90% vo ltage points at the DC output of the detector. A calibrated signal generator is needed to determine the actual DC output voltages for these input voltages.
30 28 Rec. ITU-R M.1177-4 At the measurement antenna input, insta ll a bandpass filter that Step 3: will pass the radar fundamental-frequency energy and that has bandwidth that exceeds (1/ t ) of the radar pulses to be r measured. Following the bandpass filter, install a diode detector that has bandwidth and rise time 7 t ) of the radar pulses to be measured . response speed that exceeds (1/ r Step 4: Connect the detector output to the input of an oscillosc ope. The output impedance of the detector must be matched to the input impedance of the oscilloscope. The oscilloscope must have single-shot bandwidth exceeding (1/ t ured. Set the oscilloscope to ) of the radar pulses to be meas r r threshold low enough to ensure a single-sweep mode, with a trigge that radar pulses are captured. Wait until a series of pulses are recorded. Elevat e the trigger threshold a nd wait for another set of pulses to activate the trigger. Continue this proc ess until the threshold is sufficiently high that no shold slightly, and wait for a sequence to be more pulses are recorded. Reduce the trigger thre recorded. This pulse sequence s hows the pulse repetition rate. Measure the pulse width and rise time or fall time on the oscilloscope using criteria Step 5: specified above for hard- line coupled measurements. 2.3 Notes on the radiated pulse measurement procedure By positioning the measurement system in close pr oximity to the radar, with clear line-of-sight, multipath problems are minimized and the received power in the pulses is maximized. Use of a high-gain measurement antenna further mitig ates the multipath problem and increases the received pulse power level. Care must be taken to ensure that all elements in the measurement system have bandwidth and time rise time. Diode detectors with response characteristics that are su fficient to measure radar pulse s are probably nece is requirement. fast response characteristic ssary to meet th In multi-radar environments, or environments with strong ambient non-radar signals that are in or near the edges of the spectrum band of the radar being measured, it may be necessary to take steps to isolate the pulses of the radar . The use of a parabolic antenna being measured from other signals d also a bandpass filter at th e measurement antenna terminals for microwave-frequency radars, an If these items are not adequate to isolate the will help to isolate the desired pulse waveforms. desired pulses, then amplitude-d ependent triggering should provi de the necessary isolation, assuming that pulses from the radar being measur ed have higher amplitude in the measurement system than any other signals in the environment. 3 Measurements for advanced technology radars Hardline coupled pulse measurements 3.1 In this context, advanced radars are those that ation. Either frequency or phase utilize pulse modul may be modulated. If FM (chirp ing) is being utilized, then the same measurement techniques may be used as specified above. But the measurement bandwidth must equal or exceed the total chirped- frequency range. In practice this may requi re the use of a broadband diode detector. Measurements of rise time on chirped pulses ar e the same as for non-chirped pulses; the same procedure as specified above may be used. 7 The peak input power to the detector should fall within the square-law response region. To obtain the appropriate power input level, it may be necessary to install either an attenuator or an amplifier between the bandpass filter and the diode detector.
31 Rec. ITU-R M.1177-4 29 Measurements to determine pulse compression are Pulse compression ratio (FM pulse systems): pulse compression for all radars, described below. This approach is adequate for determination of including advanced systems. width is also performed as specified above. For phase-coded radar pulses, the measurement of pulse e individual phase segments (chi ps) may be difficult. The first But measuring the rise time of th π difficulty arises with ordinary phase coding, in which a phase change of may occur between each the squared value of the wavefo rm is observed at the output of chip. Although the phase is shifted, the detector, erasing phase information. This makes the edges of the chips unobservable, detector of any sort. in principle, with a ccur at the phase transi tions between the chips, and these transients In application, transients may o on of the chip transitions does not result in are visible on an oscilloscope. But the observati a measurement of the chip rise time. Total number of subpulses within ea ch pulse (phase-coded systems): Radars employing ± 180° will normally exhibit transients conventional phase-shifting with instantaneous switching of pes of the detected pulses. In this manner, the number of chips in that can be observed in the envelo each pulse can be determined. However, for ra dar systems employing minimum shift keying (MSK) such transients, it is impossible to determine the or other phase-shifting technologies that eliminate number of chips within each pul se by measuring the detected pul se envelope. For these radars, if a pair of hardline connections to monitor the I and Q channels are not available, the number of chips can only be determined by using reference materials such as technical manuals, operating manuals and specification sheets. Chip rise time measurement: For ordinary phase-coded pulses, the chip rise time may be measured detection. This may be done by connecting an IF directly only if the waveform is sampled prior to 8 output from a spectrum analyzer to a vector signal analyzer or similar di gital signal processing device. inuous phase changes be tween chips. Instead Advanced phase-coded pulses do not employ discont they use MSK. With MSK modulati on, the requirement for observation of chip rise time is to s of the pulse, and observe th e rise time of each component separate the I and Q component individually. This can be accomplished with an appropriately programmed vector signal analyzer (VSA) (or dedicated digital signal processor (DSP) or field programmable ga te array (FPGA)) that is fed the IF output from a spectrum analyzer. If a measurement organization does not have av ailable the phase-sensitive equipment described above (VSA, DSP or FPGA with a ppropriate software installed), a pulse rise time measurement stead of a direct chip rise time measurement. may be performed on the rising edge of the pulse in The rise time measurement is performed as describe d above. If this is done, the fact should be noted in the resulting data set. 3.2 Radiated coupled pulse measurements In advanced radars that lack a directional coupler (such as systems employing multiple transmitter modules), pulse characteristics must be measured radiatively, as described above. Care must be taken to maintain adequate bandwidth for measurem ent of pulse rise time, and the diode detector input should be at an amplitude that is in the square-law response of the detector. 8 The IF output is assumed to be coupled from the spec trum analyzer prior to the detection and resolution bandwidth stages, so that adequate bandwidth is maintained for a pulse rise time measurement.
32 30 Rec. ITU-R M.1177-4 Use of reference materials to determine pulse characteristics 3.3 ferences may be presumed to be Operations manuals, specifi cation sheets and other radar-specific re reasonably accurate for the aggreg ate set of all radars within a production line or particular model series, although it must be recognized that any individual radar may vary somewhat from the ccurs as a result of bot h quality variation in production average. Such variation presumably o that the radar receives in the fi manufacturing and the maintenance eld. If one or more of the t be measured directly, the para required pulse characteristics canno meter values that are quoted in such references may be used for emission mask computations. Appendix 4 to Annex 1 Calculation of gain correction fact ors for a planar antenna array using a software progr am written in BASIC ******************************************************************************** This program is written, in BASIC, to determine the far field from a near field measurement. Uses e received wave due to only the considerations of the phase changes of th the difference between the spherical RF wavefront and the planar antenna a rray. Thus the program should only be used to at infinity from a near field measurement. determine the boresight or maximum antenna gain Antenna gain pattern is not addressed here. ******************************************************************************** 'Test data for error -.025 pi radians; error ~.3 dB 'freq = 3000 'l = 10 'd = 1 ' CLS ' INPUT “Enter the antenna frequency in MHz”; freq INPUT “Now enter th e measuring distance in metres from the antenna”; l INPUT “Enter the maximum dimensi on of the antenna in metres”; d ' ' ' CONST c = 300 CONST pi = 3.141592654# '
33 Rec. ITU-R M.1177-4 31 ' lamda = c / freq num = 100 ' ' IF d < (5 * lamda) THEN PRINT “Antenna dimensions should be much greater (* 5) than”; PRINT “the wavelength for accurate use of this prog” STOP END IF 'sum of inphase and quadrature field elements sumi = 0 sumj = 0 ' integrate from 0 to d/2 ' system is symmetrical so FOR i = 0 TO num – 1 dprime = i * d / (2 * (num – 1)) phasediff = (l – ((l ^ 2) + (dprim e ^ 2)) ^ .5) * 2 * pi / lambda ' PRINT “phase diff is”; PRINT USING “##.##”; phasediff; ' icomp = COS(phasediff) sumi = sumi + icomp jcomp = SIN(phasediff) sumj = sumj + jcomp NEXT i PRINT “Max phase error is”; PRINT USING “##.##”; phasediff / pi; PRINT “* pi radians” 'form final received planar power r eceived from spherical RF wave res = ((sumj) ^ 2 + (sumi) ^ 2) ^ .5 'PRINT “Result is”; res; “i is”; i; “num is”; num 'Calc gain reduction gprime = num / res ' glog = 20 * (LOG(gprime) / LOG(10#)) PRINT “Gain reduction from infinite far field is”;
34 32 Rec. ITU-R M.1177-4 PRINT USING “##.###”; glog; PRINT “dB” END Annex 2 Measurement of unwanted emi ssions of radar systems 2 and 3 as detailed in recommends 1 Introduction and indirect. The dire ct measurement method The techniques recommended are termed direct accurately measures unwanted emissions from radars (as detailed in recommends 2 and 3) through ated signals. The indirect me free space measurement of the radi thod measures the signals at transmitter output then combines it with models of the subsequent system to estimate the free space two techniques has shown very cl field strengths. Comparison of the ose agreement; to within 2 dB. 2 Reference bandwidth In general the rules for determining reference ba ndwidth for higher frequenc y radar (see Annex 1) apply to lower frequency radar with suitabl e scaling of the waveform parameters. For radar systems, the reference bandwidth, B , used to define unwanted emission limits ref (Recommendations ITU-R SM.329 and ITU-R SM.1541, and RR Appendix 3) should be calculated for each particular radar system. For the three gene ral types of radar pulse modulation utilized for long wavelength radionaviga tion, radiolocation, acquisition, tr acking and other radiodetermination functions, the reference bandwid th values are determined using the following formulas: – for fixed-frequency, non-pulse-coded radar, one divided by the radar pulse length, in dar pulse length is 100 seconds (e.g. if the ra s, then the reference bandwidth is μ μ s = 10 kHz); 1/100 for fixed-frequency, phase coded pulsed rada – se chip length (s) r, one divided by the pha μ s long, then the reference bandwidth is (e.g. if the phase coded chip is 200 μ s = 5 kHz); 1/200 – for FM or chirped radar, the square root of the quantity obtained by dividing the chirp 250 MHz to s) (e.g. if the FM is from 1 μ bandwidth (MHz) by the pulse length ( 1 251 MHz or 10 kHz during the pulse of 20 ms, th en the reference bandwidth is 1/2 = 700 Hz). (10 kHz/20 ms) In cases, where the above calculated bandwidths are greater than 1 MHz, then a reference bandwidth, B , of 1 MHz should be used. ref
35 Rec. ITU-R M.1177-4 33 3 Measurement bandwidth and detector parameters , is defined as the impulse bandwidth of the receiver and is greater B The measurement bandwidth, m than the IF bandwidth, on bandwidth for spectrum analyzers). B , (sometimes referred to as resoluti if B , may be derived from the following equation: The measurement bandwidth, m MBR B B × = m if The MBR needs to be determined for the measur ement receiver being used. MBR is approximately typically used in ma ny commercial spectrum 3/2 for a –3 dB IF bandwidth Gaussian filter as analyzer receivers. NOTE 1 – In some instruments the IF bandwidth is defined at the –6 dB point. to give one of the following recommended An appropriate receiver IF bandwid th should be selected l, the rules for determining measurement bandwidth for higher measurement bandwidths. (In genera frequency radar with suitable scaling of the frequency radars (see Annex 1) apply to lower waveform parameters.) ) for fixed-frequency, non- is the pulse T (1/ T Measurement pulse-coded radars, where ≤ 9 bandwidth μ s, then the measurement IF length (e.g. if radar pulse length is 100 ≤ 1/(100 μ s) = 10 kHz). bandwidth should be (1/ t is the phase- t ) for fixed-frequency, phase -coded pulsed radars, where ≤ μ s pulses, each pulse consisting o f chip length (e.g. if radar transmits 260 13 phase coded chips that are 20 μ s in length, then the measurement IF 1/(20 μ s) = 50 kHz). ≤ bandwidth should be 1/2 ( irp, or FMCW) radars, where / T ) B , for swept-frequency (FM or ch ≤ B is the range of frequenc y sweep during each pulse and T is the pulse length (e.g. if radar sweeps (chi rps) across frequency range o f 1 250-1 251 MHz ( = 10 kHz of spectrum) during each chirp, and if the chirp length is 20 ms, then the measurement IF bandwidth should be 1/2 / 20 ms) ≤ = 5 . (10 kHz kHz ≈ 700 Hz). 0 4 Dynamic range of the measurement system The measurement system should be able to meas ure levels of unwanted emissions as given in RR Appendix 3. To obtain a complete picture of the spec trum especially in the spurious emissions domain, it is recommended to be able to measure levels of emissions 10 dB below the levels given in RR Appendix 3. For a high level of confidence in the results, the measurement dynamic range of the system should be significantly higher than the required ra nge of measurement (margin (2) in Fig. 2). The link between the required ra nge of measurement and the recommended dynamic range of the measurement system is given in Fig. 2. 9 The corrections associated with measurement bandw idth transforms to reference and PEP bandwidths discussed in § 3 of Annex 1, also apply to long wavelength radars describ ed here in Annex 2.
36 34 Rec. ITU-R M.1177-4 5 Direct method A direct method, described below, can be used to measure unwante d emissions (OoB and spurious) e main beam of the radar. s, which allow easy access to th from long wavelength radar system tenna or array is situ ated on the ground and is vertically polarized. For instance where the an This method has been used to measure the em ission characteristics of long wavelength radar systems operating at frequencies up to 45 MH z, and e.i.r.p.s in the megawatt range. 5.1 Measurement hardware and software 5.1.1 Antenna A block diagram of the type of measurement syst em required for the two direct methods are shown in the system is the r in Fig. 11. The first element to be considered eceive antenna. The receive antenna should have a broadband fre as the frequency range to be quency response, at least as wide eens. Gain is not usually a problem and so measured. This may require the use of ground scr ound screen is adequate. Calibrati on of the antenna gain may be a simple whip antenna with a gr This can be achieved using a reference source and a second required for broadband measurement. short (poorly matched) antenna feeding into a power meter. The antenna should be located in th e far field if practical, for exam ple at 20 MHz, more than 1 km away, although measurement of spectral characterization has s hown no discernible difference in far field and near field measurements. Many long wa velength radars are arrays which synthesize a beam that is electronically st eerable. In this case the beam should be steered or measurement is as close as possible to antenna positioned such that the measurement antenna the peak of the main beam. The antenna polarization is selected to maximize response to the radar signal. The cable connecting the measurem ent antenna to the measurement system may be normal coaxial cable. Clear channel advisor 5.1.2 Because of ionospheric propagation long wavelength transmissions can travel large distances, and so much of the spectrum that is measured by the test antenna will, in general, be exposed to external signals. It is therefore important to have a device advising of occupied channels, preferably one that can capture this data and give some indication of the signal st rength. The spectrum measuring system can be used for this, or an independent r eceiving system. This data can be used to reconcile any unwanted emissions that may have been caused by external sources. It s hould also be used to detect a clear channel for the testing of within the in-band B and OoB domains. –40 5.1.3 RF front-end The RF front-end performs two functi ons. The first is to protect the front-end of the detection system through the use of variable RF attenuation. The s econd is low-noise preamplif ication to provide the maximum sensitivity to low power emissions. The RF attenuator is the first element in the front-end. It provides variable attenuation (e.g. 0-70 dB) in fixed increments (e.g. 10 dB/attenu ator step).
37 35 Rec. ITU-R M.1177-4 FIGURE 11 Block diagram for measurement of radiated unwanted emissions from radars using the manually controlled direct method t Measuremen antenna (whip) Radar antenna or array Aligned for maximum response in measurement system Clear channel advisor Low-loss RF line (optional) (as short as possible between antenna and measurement system input port) Measurement system RF front-end (optional) R1 or R2 LNA Variable RF attenuator used to optimize measurement system gain/noise figure trade-off Selective R1 receiver (Remote operation and data collection) Spectrum R2 analyzer Manually controlled measurement system 5.1.4 consists of sweeping across the spectrum in fixed increments The manually controlled measurement quency sweep, the attenuato (equal to the measurement bandwidth). At each fre r may be adjusted to keep the radar peak power within the dynamic range of the measurement system (often the front-end amplifier and the spectrum analyzer log amplifier are the limiting elements). With the front-end RF attenuator properly adjusted at each sw eep, a measurement of th e radar power at that frequency is performed. An LNA is installed as the next element in the The final element in the RF front-end is an LNA. signal path after the preselector. The low-nois e input characteristic of the LNA provides high sensitivity to low-amplitude spurious radar emi ssions, and its gain accommodates the noise figure of the rest of the measurement system (e.g. a length of transmissi on line and a spectrum analyzer/selected receiver).
38 36 Rec. ITU-R M.1177-4 the measurement system are op timized by proper selection of The sensitivity and dynamic range of mize noise figure while providing LNA gain and noise figure character istics. It is desirable to mini enough gain to accommodate all measurement circuitr y after the LNA (essentially the RF line loss after the front-end, plus the noise figure of the spectrum analyzer/selected receiver circuitry). NA gain and noise figure (which is the excess noise produced by the LNA Ideally, the sum of the L Ω with a 50 mately equal to the noise figure of the termination on its input) should be approxi that the spectrum an alyzer noise figure is remaining measurement system. For example, assume nd and the analyzer is 5 dB. Thus the front-end 25 dB and the RF line loss between the RF front-e LNA must accommodate a total noise figure of 30 dB. The sum of the LNA gain and noise figure should therefore be approximately 30 dB in this example. A comb ination for such an LNA would be 3 dB noise figure and 27 dB gain. The remainder of the RF measurement system is ntially a commercially expected to be esse with a pre-selector or a selective receiver. available spectrum analyzer or a spectrum analyzer the frequency range of interest, can be used. Any equipment, which can receive signals over digital receivers whic Measurements have been performed with modern h easily accommodate the frequency and dynamic range requireme nts, largely obviating the need for any attenuation or gain in the front-end. 6 Indirect method In the indirect method, measurements are made by coupling from the output of each transmitter. method from that point. If there are multiple The measurement apparatus is similar to the direct transmitters, then the complex amplitude must be recorded; then the signals must be combined together in software taking account of beam steering array weighti ng and feeder delays. Data capture can be readily achieved by connection of the spectrum analyzer or receiver to a laptop PC through a GPIB or equivalent interface.
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